Although it is also applicable to other similar semiconductor components, the present invention as well as the problems on which it is based are explained here with regard to a vertical IGBT (insulated gate bipolar transistor) for ignition applications.
IGBTs are used in general as power switching devices in the range of a blocking voltage of a few hundred volts up to a few thousand volts. In particular, the use of such IGBTs as ignition transistors, i.e., as switches on the primary side of an ignition coil, is of particular interest.
The structure of a vertical IGBT is similar to that of a VDMOS transistor, with the difference that a p+-emitter is situated on its anode side instead of an n+-substrate as in the VDMOS transistor. German Patent No. 31 10 230 describes a vertical MOSFET component having the basic structure of a vertical IGBT.
In principle, two types of vertical IGBT or V-IGBT are differentiated, namely a punch-through IGBT (PT) and a non-punch-through IGBT (NPT), as described by Laska et al., Solid-State Electronics, volume 35, no. 5, pp. 681–685, for example.
The basic properties of these two types of IGBTs are described below with reference to FIG. 6.
FIG. 6 shows a schematic cross-sectional diagram of a known PT-IGBT or NPT-IGBT, labeled in general with reference number 100.
A PT-IGBT is usually produced on a thick p+-doped substrate having an epitaxially applied n−-buffer layer 140 and an n−-drift region 104, which is also applied epitaxially. For the smallest possible conducting state voltage drop, the thickness of n−-drift region 104 is selected to be smaller than required by the width of the space charge region in n−-drift region 104 at the desired blocking ability, so n-buffer layer 140 has the function of preventing punch-through of the space charge region to rear p+-emitter 105 which is provided in the substrate. To achieve a rapid shutdown of the electric current despite a good p+-emitter 105, the carrier lifetime is kept small by lifetime killing, e.g., by radiation, and/or the doping of n-buffer layer 140 is selected to be high accordingly. Since the conducting state voltage becomes greater with an increase in the doping dose of n-buffer layer 140, a good compromise between conducting-state behavior and shutdown behavior is to be achieved with a thin, highly doped n-buffer layer 140.
An NPT-IGBT is derivable from the PT-IGBT by omitting n-buffer layer 140 and selecting a greater thickness for drift region 104 than that required by the width of the space charge region at the desired blocking ability. The NPT-IGBT is usually produced on a low-doped substrate having a high charge carrier lifetime, whereby after introducing the diffusion profiles on the front side of the wafer, a shallow p+-emitter 105 having a depth of penetration of only a few μm (much less than 20 μm) and a poor emitter efficiency is produced on the rear side of the wafer. Such a transparent p+-emitter 105 is used to ensure a rapid shutdown of the current in dynamic operation of the component with the goal of minimizing shutdown losses. To achieve satisfactory conducting state properties despite the poor p+-emitter 105, the carrier lifetime in n−-drift region 104 must be selected to be as high as possible, and furthermore, the thickness of n−-drift region 104 is to be selected to be as low as possible, taking into account the desired blocking ability of the component.
On the front side, a PT-IGBT or NPT-IGBT is composed of an active region 130 and an edge termination region 150, the latter ensuring the desired blocking ability with respect to the edge of the chip. Active region 130 is composed of a plurality of cell-shaped or strip-shaped MOS control heads 106, 107, 108 connected in parallel. These MOS control heads 106, 107, 108 are explained in greater detail below in conjunction with the functioning of vertical IGBTs.
MOS control heads 106, 107, 108 are obtained by continued reflection of the half-cell on section AA′ shown between sections AA′ and BB′ in FIG. 6. Magnetoresistive structures are conventionally used in edge area 150 to achieve the desired blocking ability. These magnetoresistive structures are usually composed of a cathode 101, designed as a metal magnetoresistor, a polysilicon magnetoresistor 153a, which is electrically connected to the former via the third dimension (not shown), a metal plating 152 connected to an n+-channel stopper 155 and a polysilicon magnetoresistor 153b, which is electrically connected to metal plating 152 via the third dimension (not shown). Also shown are a field oxide 159 and an intermediate dielectric 110, which, in addition to specific contacting, has the function of electrically insulating the metal plating plane from the polysilicon plane.
The functioning of an NPT-IGBT or PT-IGBT in the conducting state is explained first in greater detail below.
A gate 103, which is usually made of polysilicon and is insulated from the semiconductor body by a thin gate oxide layer 109, is brought to a potential above the threshold voltage of MOS control heads 106, 107, 108 with respect to cathode 101. Then an inversion channel is created on the semiconductor surface beneath gate 103 in the area of p-body region 108, whereupon the semiconductor surface is in a state of accumulation in the area of n-drift region 104. When there is a positive voltage on anode 102 relative to cathode 101, electrons are injected into n−-drift region 104 via n+-source region 106, the MOS channel thus influenced, and the accumulation layer. Then p+-emitter 105 on the anode side injects holes, so that n−-drift region 104 is flooded by charge carriers and its conductivity is increased in active region 130 and in adjacent portions of edge termination 150. These portions are in high injection at the usual conducting state current densities. Therefore, an IGBT having a blocking ability of more than approx. 150–200 V is capable of carrying a higher current density with a smaller voltage drop between the anode and cathode than a MOS transistor having the same breakdown voltage. In the conducting state, current flows from anode 102 to cathode 101. It is carried by electrons, which are injected into n−-drift region 104 and flow out over p+-emitter 105 on the anode side to anode 102, and it is also carried by holes, which are injected from the p+-emitter on the anode side into n−-drift region 104 and flow over p-regions 107, 108 to cathode 101.
In addition to the planar vertical IGBT structures discussed here, there are also vertical IGBTs having a trench gate, where the gate is created in the form of a trench in the semiconductor surface. In this regard, see I. Omura et al., ISPSD '97, Conf. Proc., pp. 217–200. The functioning of these vertical IGBTs having a trench gate is completely similar to that of the structures described above, but they offer the advantage of a lower conducting state voltage drop.
The functioning of the NPT-IGBT or PT-IGBT in the blocking case will be described now. In the blocking case, gate 103 is brought to a voltage below the threshold voltage relative to cathode 101. If anode 102 is then brought to a positive potential, the space charge region, which has developed at the boundary between p-body region 108 and n−-drift region 104, expands almost exclusively into n−-drift region 104.
In the NPT-IGBT, the thickness of n−-drift zone 104 is greater than the width of the space charge region at a given maximum blocking ability of the component. This results in the triangular curve (dotted line) of electric field strength |E| along thickness direction y of the component, as indicated in FIG. 6. The maximum of field strength |E| occurs in the area of MOS control heads 106, 107, 108.
In the PT-IGBT, the thickness of n−-drift zone 104 is selected to be less than the width the space charge region would have at a given maximum blocking ability of the component. To prevent the space charge region from running up onto rear p+-emitter 105, n-doped buffer layer 140 is introduced here with the goal of preventing punch-through. This results in the trapezoidal curve (solid line) of electric field strength |E| indicated in FIG. 6 along thickness direction y of the component. The maximum field strength also occurs here in the area of MOS control heads 106, 107, 108.
FIG. 7 shows a conventional circuit topology in which a vertical IGBT 100 according to FIG. 6 is used as the ignition transistor in the primary circuit of an ignition coil for an internal combustion engine. For this use as an ignition transistor, a V-IGBT having a required blocking ability of approx. 400–600 V has been used in the past.
According to FIG. 7, V-IGBT 100 having main terminal 101 corresponding to the cathode, main terminal 102 corresponding to the anode and control terminal 103 corresponding to the gate is connected to battery voltage VBat at node 210 via a primary winding of an ignition coil 211. A spark plug 212, a protective resistor 214 of 1–2 kΩ and a diode 213 for suppressing the starting spark are connected on the secondary winding side of ignition coil 211.
V-IGBT 100 is integrated into a circuit system 200 having terminal nodes 201, 202 and 203. Terminal node 202 is connected directly to first main terminal 102 of V-IGBT 100, and terminal node 203, which is at ground GND, is connected directly to second main terminal 101 of V-IGBT 100.
The additional circuit components within circuit system 200 have the function of triggering and clamping V-IGBT 100. Diode 204 has the function of protecting gate 103, which is connected to it, from overvoltages. When conducting, diode 206 prevents current from flowing from control terminal 103 to main terminal 102, which is connected to terminal 152 via the semiconductor material of the V-IGBT. Resistors 207 of 1 kΩ, for example, and resistors 208 of 10–25 kΩ, for example, determine the input resistance of circuit system 200 at terminal node 201 for a control signal ST, and they also form the load of a clamping diode device 205, which is usually designed as a clamping diode chain having a plurality of polysilicon Zener diodes polarized in the reverse direction. Elements 204, 205, 206, 207 and 208 are usually integrated monolithically with the V-IGBT, diodes 204, 206, in addition to element 205, being also normally made of polysilicon.
As outlined here, clamping diode device 205 is not connected directly to the metal plating of anode 102 because it is on the bottom side of the chip and is not readily accessible. Instead, it is in contact with metal plating 152 of channel stopper 155 which has the same potential as anode 102 up to a forward voltage. Circuit system 200 is directly operable by a control unit via terminal node 201. To do so, a control signal ST having a positive voltage of 5 V, for example, is applied to terminal node 201, whereupon a current rise is initiated by ignition coil 211.
At a certain point in time, the voltage on terminal node 201 is reduced to approx. 0 V, whereupon the voltage on metal plating 152 and on main terminal 102 and thus on terminal node 202 rises steeply. The voltage rise is transformed up on the secondary side of ignition coil 211, resulting in an ignition spark on spark plug 212. Clamping diode chain 205 has the function of limiting the voltage rise at main terminal 102 to so-called clamp voltage VKL of approx. 400 V to protect V-IGBT 100 and the other circuit components of circuit system 200. This is important in the pulse case in particular.
A pulse case occurs when no ignition spark is generated, e.g., due to a detached ignition cable. Then circuit system 200 including V-IGBT 100 must absorb the energy which would otherwise be converted into a spark.
FIG. 3 shows a schematic diagram of the time characteristic of the clamping of the anode voltage with conventional circuit system 200.
A time characteristic of voltage VA at first main terminal 102, i.e., on terminal node 202 is shown in FIG. 3 with dotted line 302. It is assumed here that the ignition switch was switched on for a certain period of time during t<0, so that at time t=0, a current of typically 7 to 20 A is flowing through V-IGBT 100 and ignition coil 211. If V-IGBT 100 is switched off at t=0 due to the reduction in the voltage of triggering signal ST at terminal node 201 to 0 V, then ignition coil 211 is still forcing the full current on it at first.
Then voltage VA at first main terminal 102 rises steeply. Without voltage limiting, voltage VA at first main terminal 102 would then rise to the breakdown value of V-IGBT 100 and would destroy it. This is prevented by clamping diode device 205 in that on reaching preselected clamp voltage VKL at time tr (tr typically amounts to a few μs), gate 103 of V-IGBT 100 is triggered so sharply that it prevents clamp voltage VKL at main terminal 102 from being exceeded.
If the case is not a pulse case but instead a standard operating case according to solid-line curve 301 in FIG. 3, then voltage VA at first main terminal 102 would collapse after approx. t1–tr=15 μs, and after another approx. 15 μs, it would generate the ignition spark on spark plug 212 at tf. The result would be a conversion of the energy stored in ignition coil 211 during sparking period t3–tf in the combustion chamber, during which only the inversely transformed spark voltage of approx. VB=30 V would be applied to first main terminal 102 for most of the time period at the end of sparking period t3–tf voltage VA at main terminal 102 would again drop to battery voltage VBat=14 V.
In the pulse case, represented by dotted-line curve 302 in FIG. 3, however, the high clamp voltage of approx. 400 V would persist until time t4, and the current flowing through ignition coil 211 and V-IGBT 100 would subsequently drop linearly over time until time t4. At time t4, the coil energy will have dissipated, i.e., been converted to heat in circuit system 200, and voltage VA at terminal 102 will drop steeply to battery voltage VBat. Period of time t4–tr lasts only a few hundred μs, but this operating case nevertheless makes a high demand on the pulse strength of IGBT 100 due to the high power converted, and this requirement is not always met to an adequate extent. In the worst case this results in destruction of IGBT 100.
J. Yedinak et al. (ISPSD '98, Conf. Proc., pp. 399–402) show, using the example of a PT-IGBT that failure occurs as follows. In the pulse case, the space charge region has included the entire n−-drift region 104. Electrons are injected via the MOS channel that has developed into the n−-drift region 104 via a triggering of gate 103 controlled by clamping diode 205, these electrons triggering rear p+-emitter 105. The component becomes very hot, in particular in the area of cathode 101, due to the high current density, the high electric field strength and the associated high power loss in the area of MOS control heads 106, 107, 108, whereupon an electron leakage current occurs from MOS control heads 106, 107, 108. The electrons run toward anode 102 and notch up p+-emitter 105. They thus function like additional triggering of IGBT 100.
To keep the voltage at the level of the clamp voltage, the triggering of gate 103 is reduced accordingly via clamping diode chain 205. Under certain operating conditions, the notch-up due to the thermally induced electron leakage current is so great that V-IGBT 100 is able to carry the load current without gate notch-up. Its controllability is lost. Then the temperature and leakage current of the component continue to rise. Finally, thermal direct feedback occurs and V-IGBT 100 is destroyed. In an investigation of the dependence of the pulse strength of V-IGBTs on the clamp voltage according to Z. J. Shen et al., IEEE Electron Device Letters, volume 21, no. 3, March 2000, pp. 119–122, it was found that the pulse strength increases greatly with a decrease in clamp voltage. The reason for this is the reduction in the power converted in V-IGBT 100 due to the drop in clamp voltage; thereby the maximum temperature which occurs in the area of MOS control heads 106, 107, 108 drops during a pulse case.
If we consider standard ignition systems in a motor vehicle, we find that the clamp voltage in such a system is not freely selectable and in particular is not significantly reducible. A significantly reduced clamp voltage would endanger reliable generation of the ignition spark.
An intelligent V-IGBT having current limiting and an over-temperature cutoff is described by Z. J. Shen et al., PCIM '96, Conf. Proc., pp. 11–16, where polysilicon diodes are used as the temperature sensor. In the on-state, the IGBT is shut down on reaching a certain threshold temperature due to the fact that the monolithically integrated control circuit reduces the gate voltage. However, this IGBT is not suitable for use as an ignition transistor because it does not have any clamping. In addition, an overtemperature cutoff by reducing the gate voltage would be counterproductive in the pulse case because it would render ineffective the clamping, which also intervenes via the gate voltage.
An object of the present invention is thus to create an improved semiconductor circuit configuration, in particular for ignition applications, having a semiconductor power switching device which is more protectable in a pulse case.